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LT1373 250kHz Low Supply Current High Efficiency 1.5A Switching Regulator
FEATURES
s s s s s s s s s s s
DESCRIPTIO
1mA IQ at 250kHz Uses Small Inductors: 15H All Surface Mount Components Only 0.6 Square Inch of Board Space Low Minimum Supply Voltage: 2.7V Constant Frequency Current Mode Current Limited Power Switch: 1.5A Regulates Positive or Negative Outputs Shutdown Supply Current: 12A Typ Easy External Synchronization 8-Pin SO or PDIP Packages
The LT (R)1373 is a low supply current high frequency current mode switching regulator. It can be operated in all standard switching configurations including boost, buck, flyback, forward, inverting and "Cuk." A 1.5A high efficiency switch is included on the die, along with all oscillator, control and protection circuitry. All functions of the LT1373 are integrated into 8-pin SO/PDIP packages. Compared to the 500kHz LT1372, which draws 4mA of quiescent current, the LT1373 switches at 250kHz, typically consumes only 1mA and has higher efficiency. High frequency switching allows for small inductors to be used. All surface mount components consume less than 0.6 square inch of board space. New design techniques increase flexibility and maintain ease of use. Switching is easily synchronized to an external logic level source. A logic low on the shutdown pin reduces supply current to 12A. Unique error amplifier circuitry can regulate positive or negative output voltage while maintaining simple frequency compensation techniques. Nonlinear error amplifier transconductance reduces output overshoot on start-up or overload recovery. Oscillator frequency shifting protects external components during overload conditions.
, LTC and LT are registered trademarks of Linear Technology Corporation.
APPLICATIO S
s s s s s
Boost Regulators CCFL Backlight Driver Laptop Computer Supplies Multiple Output Flyback Supplies Inverting Supplies
TYPICAL APPLICATIO
5V 5 OFF ON 4 S/S
VIN
5V-to-12V Boost Converter
L1* 22H D1 MBRS120T3 R1 215k 1% VOUT 12V
EFFICIENCY (%)
100 VIN = 5V f = 250kHz 90
VSW
LT1373
8
80
+
+
C1** 22F
FB GND 6, 7 VC 1
2
C4** 22F MAX IOUT
70
C2 0.01F R3 5k
R2 24.9k 1%
L1 IOUT 15H 0.3A 22H 0.35A *SUMIDA CD75-220KC (22H) OR COILCRAFT D03316-153 (15H) **AVX TPSD226M025R0200
60
50 1 10 100 OUTPUT CURRENT (mA) 1000
LT1373 * TA02
LT1373 * TA01
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12V Output Efficiency
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LT1373
ABSOLUTE
(Note 1)
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RATI GS
PACKAGE/ORDER I FOR ATIO
TOP VIEW VC 1 FB 2 NFB 3 S/S 4 N8 PACKAGE 8-LEAD PDIP 8 7 6 5 VSW GND GND S VIN
Supply Voltage ....................................................... 30V Switch Voltage LT1373 ............................................................... 35V LT1373HV .......................................................... 42V S/S Pin Voltage ....................................................... 30V Feedback Pin Voltage (Transient, 10ms) .............. 10V Feedback Pin Current ........................................... 10mA Negative Feedback Pin Voltage (Transient, 10ms) ............................................. 10V Operating Junction Temperature Range Commercial ........................................ 0C to 125C* Industrial ......................................... - 40C to 125C Short Circuit ......................................... 0C to 150C Storage Temperature Range ................ - 65C to 150C Lead Temperature (Soldering, 10 sec)................. 300C
ORDER PART NUMBER LT1373CN8 LT1373HVCN8 LT1373CS8 LT1373HVCS8 LT1373IN8 LT1373HVIN8 LT1373IS8 LT1373HVIS8
S8 PACKAGE 8-LEAD PLASTIC SO
TJMAX = 125C, JA = 100C/ W (N8) TJMAX = 125C, JA = 120C/ W (S8)
S8 PART MARKING 1373 1373I 1373H 1373HI
Consult factory for Military grade parts. *Units shipped prior to Date Code 9552 are rated at 100C maximum operating temperature.
ELECTRICAL CHARACTERISTICS
The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.
SYMBOL VREF IFB PARAMETER Reference Voltage Feedback Input Current Reference Voltage Line Regulation VNFB INFB Negative Feedback Reference Voltage Negative Feedback Input Current Negative Feedback Reference Voltage Line Regulation gm Error Amplifier Transconductance Error Amplifier Source Current Error Amplifier Sink Current Error Amplifier Clamp Voltage AV f Error Amplifier Voltage Gain VC Pin Threshold Switching Frequency Duty Cycle = 0% 2.7V VIN 25V 0C TJ 125C - 40C TJ 0C (I Grade)
q q
CONDITIONS Measured at Feedback Pin VC = 0.8V VFB = VREF
q q
MIN 1.230 1.225
TYP 1.245 1.245 50
MAX 1.260 1.265 150 275 0.03 - 2.39 - 2.35 -2 0.05 500 600 90 1500 2.30 0.52 1.25 275 290 290 500
UNITS V V nA nA %/V V V A %/V mho mho A A V V V/ V V kHz kHz kHz % ns
2.7V VIN 25V, VC = 0.8V Measured at Negative Feedback Pin Feedback Pin Open, VC = 0.8V VNFB = VNFR 2.7V VIN 25V, VC = 0.8V IC = 5A
q q q q
0.01 - 2.51 - 2.55 - 12 - 2.45 - 2.45 -7 0.01 250 150 25 1.70 0.25 0.8 225 210 200 90 375 50 850 1.95 0.40 250 1 250 250 95 340
q
VFB = VREF - 150mV, VC = 1.5V VFB = VREF + 150mV, VC = 1.5V High Clamp, VFB = 1V Low Clamp, VFB = 1.5V
q q
Maximum Switch Duty Cycle Switch Current Limit Blanking Time
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LT1373
ELECTRICAL CHARACTERISTICS
The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.
SYMBOL BV PARAMETER Output Switch Breakdown Voltage CONDITIONS LT1373 LT1373HV 0C TJ 125C - 40C TJ 0C (I Grade) ISW = 1A Duty Cycle = 50% Duty Cycle = 80% (Note 2)
q q q
MIN 35 42 40
TYP 47 47 0.5
MAX
UNITS V V V
VSAT ILIM IIN ISW
Output Switch "On" Resistance Switch Current Limit Supply Current Increase During Switch On-Time Control Voltage to Switch Current Transconductance Minimum Input Voltage
0.85 2.7 2.5 20
A A mA/A A/V
q q
1.5 1.3
1.9 1.7 10 2
q
2.4 1 12 0.6 5 - 10 300 1.3 12
2.7 1.5 30 50 2 100 15 340
V mA A A V s A kHz
IQ
Supply Current Shutdown Supply Current
2.7V VIN 25V 2.7V VIN 25V, VS/S 0.6V 0C TJ 125C - 40C TJ 0C (I Grade) 2.7V VIN 25V 0V VS/S 5V
q q q q q q
Shutdown Threshold Shutdown Delay S/S Pin Input Current Synchronization Frequency Range
Note 1: Absolute Maximum Ratings are those values beyond which the life of the device may be impaired.
Note 2: For duty cycles (DC) between 50% and 90%, minimum guaranteed switch current is given by ILIM = 0.667 (2.75 - DC).
TYPICAL PERFOR A CE CHARACTERISTICS
Switch Saturation Voltage vs Switch Current
1.0
SWITCH SATURATION VOLTAGE (V)
0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 0
150C 100C
SWITCH CURRENT LIMIT (A)
INPUT VOLTAGE (V)
-55C
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 SWITCH CURRENT (A)
LT1373 * G01
UW
25C
Switch Current Limit vs Duty Cycle
3.0 2.5 2.0 -55C 1.5 1.0 0.5 0 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%)
LT1373 * G02
Minimum Input Voltage vs Temperature
3.0 2.8
25C AND 125C
2.6 2.4 2.2 2.0 1.8 -50 -25
0
25 50 75 100 125 150 TEMPERATURE (C)
LT1373 * G03
3
LT1373 TYPICAL PERFOR A CE CHARACTERISTICS
Shutdown Delay and Threshold vs Temperature
MINIMUM SYNCHRONIZATION VOLTAGE (VP-P)
20 18 16
1.8 SHUTDOWN THRESHOLD 1.6 1.4 1.2 1.0 SHUTDOWN DELAY 0.8 0.6 0.4 0.2 0 0 25 50 75 100 125 150 TEMPERATURE (C)
LT1373 * G04
fSYNC = 330kHz
ERROR AMPLIFIER OUTPUT CURRENT (A)
SHUTDOWN DELAY (s)
14 12 10 8 6 4 2 0 -50 -25
S/S Pin Input Current vs Voltage
5 4
SWITCHING FREQUENCY (% OF TYPICAL)
VIN = 5V
TRANSCONDUCTANCE (mho)
S/S PIN INPUT CURRENT (A)
3 2 1 0 -1 -2 -3 -4 -5 -1 0 1 23456 S/S PIN VOLTAGE (V) 7 8 9
VC Pin Threshold and High Clamp Voltage vs Temperature
2.4 2.2 2.0 VC HIGH CLAMP
FEEDBACK INPUT CURRENT (nA)
VFB = VREF 350 300 250 200 150 100 50 0 25 50 75 100 125 150 TEMPERATURE (C)
LT1373 * G10
NEGATIVE FEEDBACK INPUT CURRENT (A)
VC PIN VOLTAGE (V)
1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 -50 -25 VC THRESHOLD
4
UW
LT1373 * G07
Minimum Synchronization Voltage vs Temperature
2.0
Error Amplifier Output Current vs Feedback Pin Voltage
100 75 50 25 0 -25 -50 -75 -0.3 VREF -0.2 -0.1 FEEDBACK PIN VOLTAGE (V) 0.1
LT1373 * G06
3.0 2.5 2.0 1.5 1.0 0.5
25C -55C 125C
SHUTDOWN THRESHOLD (V)
0 -50 -25
0
25 50 75 100 125 150 TEMPERATURE (C)
LT1373 * G05
Switching Frequency vs Feedback Pin Voltage
110 100 90 80 70 60 50 40 30 20 10 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 FEEDBACK PIN VOLTAGE (V)
LT1373 * G08
Error Amplifier Transconductance vs Temperature
500 gm = 400 I (VC) V (FB)
300
200
100
0 -50 -25
0
25 50 75 100 125 150 TEMPERATURE (C)
LT1373 * G09
Feedback Input Current vs Temperature
400
Negative Feedback Input Current vs Temperature
0 -2 -4 -6 -8 -10 -12 -14 -16 -18 -20 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C)
LT1373 * G12
VNFB = VNFR
0 -50 -25
0
25 50 75 100 125 150 TEMPERATURE (C)
LT1373 * G11
LT1373
PI FU CTIO S
VC (Pin 1): Compensation Pin. The VC pin is used for frequency compensation, current limiting and soft start. It is the output of the error amplifier and the input of the current comparator. Loop frequency compensation can be performed with an RC network connected from the VC pin to ground. FB (Pin 2): The feedback pin is used for positive output voltage sensing and oscillator frequency shifting. It is the inverting input to the error amplifier. The noninverting input of this amplifier is internally tied to a 1.245V reference. Load on the FB pin should not exceed 100A when the NFB pin is used. See Applications Information. NFB (Pin 3): The negative feedback pin is used for negative output voltage sensing. It is connected to the inverting input of the negative feedback amplifier through a 400k source resistor. S/S (Pin 4): Shutdown and Synchronization Pin. The S/S pin is logic level compatible. Shutdown is active low and the shutdown threshold is typically 1.3V. For normal operation, pull the S/S pin high, tie it to VIN or leave it floating. To synchronize switching, drive the S/S pin between 300kHz and 340kHz. VIN (Pin 5): Input Supply Pin. Bypass VIN with 10F or more. The part goes into undervoltage lockout when VIN drops below 2.5V. Undervoltage lockout stops switching and pulls the VC pin low. GND S (Pin 6): The ground sense pin is a "clean" ground. The internal reference, error amplifier and negative feedback amplifier are referred to the ground sense pin. Connect it to ground. Keep the ground path connection to the output resistor divider and the VC compensation network free of large ground currents. GND (Pin 7): The ground pin is the emitter connection of the power switch and has large currents flowing through it. It should be connected directly to a good quality ground plane. VSW (Pin 8): The switch pin is the collector of the power switch and has large currents flowing through it. Keep the traces to the switching components as short as possible to minimize radiation and voltage spikes.
BLOCK DIAGRA
S/S
400k NFB 200k
FB
1.245V REF GND SENSE
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VIN SHUTDOWN DELAY AND RESET 250kHz OSC 5:1 FREQUENCY SHIFT LOW DROPOUT 2.3V REG
SW
ANTI-SAT
SYNC
LOGIC
DRIVER
SWITCH
+
NEGATIVE FEEDBACK AMP
-
COMP
+
CURRENT AMP VC AV 6 0.08
ERROR AMP
-
GND
LT1373 * BD
5
LT1373
OPERATIO
The LT1373 is a current mode switcher. This means that switch duty cycle is directly controlled by switch current rather than by output voltage. Referring to the Block Diagram, the switch is turned "On" at the start of each oscillator cycle. It is turned "Off" when switch current reaches a predetermined level. Control of output voltage is obtained by using the output of a voltage sensing error amplifier to set current trip level. This technique has several advantages. First, it has immediate response to input voltage variations, unlike voltage mode switchers which have notoriously poor line transient response. Second, it reduces the 90 phase shift at mid-frequencies in the energy storage inductor. This greatly simplifies closed-loop frequency compensation under widely varying input voltage or output load conditions. Finally, it allows simple pulse-by-pulse current limiting to provide maximum switch protection under output overload or short conditions. A low dropout internal regulator provides a 2.3V supply for all internal circuitry. This low dropout design allows input voltage to vary from 2.7V to 25V with virtually no change in device performance. A 250kHz oscillator is the basic clock for all internal timing. It turns "On" the output switch via the logic and driver circuitry. Special adaptive anti-sat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. This minimizes driver dissipation and provides very rapid turn-off of the switch. A 1.245V bandgap reference biases the positive input of the error amplifier. The negative input of the amplifier is brought out for positive output voltage sensing. The error amplifier has nonlinear transconductance to reduce out-
APPLICATIO S I FOR ATIO
Positive Output Voltage Setting
The LT1373 develops a 1.245V reference (VREF) from the FB pin to ground. Output voltage is set by connecting the FB pin to an output resistor divider (Figure 1). The FB pin bias current represents a small error and can usually be ignored for values of R2 up to 25k. The suggested value for R2 is 24.9k. The NFB pin is normally left open for positive output applications.
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put overshoot on start-up or overload recovery. When the feedback voltage exceeds the reference by 40mV, error amplifier transconductance increases ten times, which reduces output overshoot. The feedback input also invokes oscillator frequency shifting, which helps protect components during overload conditions. When the feedback voltage drops below 0.6V, the oscillator frequency is reduced 5:1. Lower switching frequency allows full control of switch current limit by reducing minimum switch duty cycle. Unique error amplifier circuitry allows the LT1373 to directly regulate negative output voltages. The negative feedback amplifier's 400k source resistor is brought out for negative output voltage sensing. The NFB pin regulates at - 2.45V while the amplifier output internally drives the FB pin to 1.245V. This architecture, which uses the same main error amplifier, prevents duplicating functions and maintains ease of use. (Consult Linear Technology Marketing for units that can regulate down to - 1.25V.) The error signal developed at the amplifier output is brought out externally. This pin (VC) has three different functions. It is used for frequency compensation, current limit adjustment and soft starting. During normal regulator operation this pin sits at a voltage between 1V (low output current) and 1.9V (high output current). The error amplifier is a current output (gm) type, so this voltage can be externally clamped for lowering current limit. Likewise, a capacitor coupled external clamp will provide soft start. Switch duty cycle goes to zero if the VC pin is pulled below the control pin threshold, placing the LT1373 in an idle mode.
VOUT R1 FB PIN R2 VREF
LT1373 * F01
VOUT = VREF 1 + R1 R2 R1 = R2 VOUT -1 1.245
() ()
Figure 1. Positive Output Resistor Divider
LT1373
APPLICATIO S I FOR ATIO
Negative Output Voltage Setting
The LT1373 develops a - 2.45V reference (VNFR) from the NFB pin to ground. Output voltage is set by connecting the NFB pin to an output resistor divider (Figure 2). The - 7A NFB pin bias current (INFB) can cause output voltage errors and should not be ignored. This has been accounted for in the formula in Figure 2. The suggested value for R2 is 2.49k. The FB pin is normally left open for negative output applications. See Dual Polarity Output Voltage Sensing for limitations of FB pin loading when using the NFB pin.
-VOUT INFB NFB PIN R2 VNFR R1 -VOUT = VNFB 1 + R1 + INFB (R1) R2 R1 = 2.45 + (7 * 10 -6) R2
LT1373 * F02
()
()
VOUT - 2.45
Figure 2. Negative Output Resistor Divider
Dual Polarity Output Voltage Sensing Certain applications benefit from sensing both positive and negative output voltages. One example is the Dual Output Flyback Converter with Overvoltage Protection circuit shown in the Typical Applications section. Each output voltage resistor divider is individually set as described above. When both the FB and NFB pins are used, the LT1373 acts to prevent either output from going beyond its set output voltage. For example in this application, if the positive output were more heavily loaded than the negative, the negative output would be greater and would regulate at the desired set-point voltage. The positive output would sag slightly below its set-point voltage. This technique prevents either output from going unregulated high at no load. Please note that the load on the FB pin should not exceed 100A when the NFB pin is used. This situation occurs when the resistor dividers are used at both FB and NFB. True load on FB is not the full divider current unless the positive output is shorted to ground. See Dual Output Flyback Converter application. Shutdown and Synchronization The dual function S/S pin provides easy shutdown and synchronization. It is logic level compatible and can be pulled high, tied to VIN or left floating for normal operation.
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A logic low on the S/S pin activates shutdown, reducing the part's supply current to 12A. Typical synchronization range is from 1.05 and 1.8 times the part's natural switching frequency, but is only guaranteed between 300kHz and 340kHz. A 12s resetable shutdown delay network guarantees the part will not go into shutdown while receiving a synchronization signal. Caution should be used when synchronizing above 330kHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. This type of subharmonic switching only occurs when the duty cycle of the switch is above 50%. Higher inductor values will tend to eliminate problems. Thermal Considerations Care should be taken to ensure that the worst-case input voltage and load current conditions do not cause excessive die temperatures. The packages are rated at 120C/W for SO (S8) and 130C/W for PDIP (N8). Average supply current (including driver current) is: IIN = 1mA + DC (ISW/60 + ISW * 0.004) ISW = switch current DC = switch duty cycle Switch power dissipation is given by: PSW = (ISW)2 * RSW * DC RSW = output switch "On" resistance Total power dissipation of the die is the sum of supply current times supply voltage plus switch power: PD(TOTAL) = (IIN * VIN) + PSW Choosing the Inductor For most applications the inductor will fall in the range of 10H to 50H. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output current because they reduce peak current seen by the power switch which has a 1.5A limit. Higher values also reduce input ripple voltage, and reduce core loss. When choosing an inductor you might have to consider maximum load current, core and copper losses, allowable
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LT1373
APPLICATIO S I FOR ATIO
component height, output voltage ripple, EMI, fault current in the inductor, saturation, and of course, cost. The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. Assume that the average inductor current (for a boost converter) is equal to load current times VOUT/VIN and decide whether or not the inductor must withstand continuous overload conditions. If average inductor current at maximum load current is 0.5A, for instance, a 0.5A inductor may not survive a continuous 1.5A overload condition. Also, be aware that boost converters are not short-circuit protected, and that under output short conditions, inductor current is limited only by the available current of the input supply. 2. Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, especially with smaller inductors and lighter loads, so don't omit this step. Powered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core materials fall in between somewhere. The following formula assumes continuous mode operation, but it errors only slightly on the high side for discontinuous mode, so it can be used for all conditions. IPEAK = IOUT * VOUT VIN (VOUT - VIN) + VIN 2(f)(L)(VOUT)
VIN = minimum input voltage f = 250kHz switching frequency 3. Decide if the design can tolerate an "open" core geometry like a rod or barrel, which have high magnetic field radiation, or whether it needs a closed core like a toroid to prevent EMI problems. One would not want an open core next to a magnetic storage media for instance! This is a tough decision because the rods or barrels are temptingly cheap and small, and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. 4. Start shopping for an inductor which meets the requirements of core shape, peak current (to avoid saturation), average current (to limit heating), and fault current, (if the
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inductor gets too hot, wire insulation will melt and cause turn-to-turn shorts). Keep in mind that all good things like high efficiency, low profile and high temperature operation will increase cost, sometimes dramatically. 5. After making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. Use the experts in the Linear Technology application department if you feel uncertain about the final choice. They have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. Output Capacitor The output capacitor is normally chosen by its effective series resistance (ESR), because this is what determines output ripple voltage. At 500kHz, any polarized capacitor is essentially resistive. To get low ESR takes volume, so physically smaller capacitors have high ESR. The ESR range for typical LT1373 applications is 0.05 to 0.5. A typical output capacitor is an AVX type TPS, 22F at 25V, with a guaranteed ESR less than 0.2. This is a "D" size surface mount solid tantalum capacitor. TPS capacitors are specially constructed and tested for low ESR, so they give the lowest ESR for a given volume. To further reduce ESR, multiple output capacitors can be used in parallel. The value in microfarads is not particularly critical and values from 22F to greater than 500F work well, but you cannot cheat mother nature on ESR. If you find a tiny 22F solid tantalum capacitor, it will have high ESR and output ripple voltage will be terrible. Table 1 shows some typical solid tantalum surface mount capacitors.
Table 1. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current
E CASE SIZE AVX TPS, Sprague 593D AVX TAJ D CASE SIZE AVX TPS, Sprague 593D AVX TAJ C CASE SIZE AVX TPS AVX TAJ B CASE SIZE AVX TAJ 2.5 to 10 0.16 to 0.08 0.2 (Typ) 1.8 to 3.0 0.5 (Typ) 0.22 to 0.17 0.1 to 0.3 0.9 to 2.0 0.7 to 1.1 0.36 to 0.24 ESR (MAX ) 0.1 to 0.3 0.7 to 0.9 RIPPLE CURRENT (A) 0.7 to 1.1 0.4
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LT1373
APPLICATIO S I FOR ATIO
Many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents. This is historically true and type TPS capacitors are specially tested for surge capability, but surge ruggedness is not a critical issue with the output capacitor. Solid tantalum capacitors fail during very high turn-on surges, which do not occur at the output of regulators. High discharge surges, such as when the regulator output is dead shorted, do not harm the capacitors. Single inductor boost regulators have large RMS ripple current in the output capacitor, which must be rated to handle the current. The formula to calculate this is: Output Capacitor Ripple Current (RMS) DC IRIPPLE (RMS) = IOUT 1 - DC = IOUT Input Capacitors The input capacitor of a boost converter is less critical due to the fact that the input current waveform is triangular, and does not contain large squarewave currents as is found in the output capacitor. Capacitors in the range of 10F to 100F with an ESR (effective series resistance) of 0.3 or less work well up to a full 1.5A switch current. Higher ESR capacitors may be acceptable at low switch currents. Input capacitor ripple current for boost converter is: IRIPPLE = 0.3(VIN)(VOUT - VIN) (f)(L)(VOUT) VOUT - VIN VIN
f = 250kHz switching frequency The input capacitor can see a very high surge current when a battery or high capacitance source is connected "live", and solid tantalum capacitors can fail under this condition. Several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (AVX TPS series, for instance), but even these units may fail if the input voltage approaches the maximum voltage rating of the capacitor. AVX recommends derating capacitor voltage by 2:1 for high surge applications. Ceramic and
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aluminum electrolytic capacitors may also be used and have a high tolerance to turn-on surges. Ceramic Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low ESR. Unfortunately, the ESR is so low that it can cause loop stability problems. Solid tantalum capacitor ESR generates a loop "zero" at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually resonate with their ESL before ESR becomes effective. They are appropriate for input bypassing because of their high ripple current ratings and tolerance of turn-on surges. Linear Technology plans to issue a Design Note on the use of ceramic capacitors in the near future. Output Diode The suggested output diode (D1) is a 1N5818 Schottky or its Motorola equivalent, MBR130. It is rated at 1A average forward current and 30V reverse voltage. Typical forward voltage is 0.42V at 1A. The diode conducts current only during switch-off time. Peak reverse voltage for boost converters is equal to regulator output voltage. Average forward current in normal operation is equal to output current. Frequency Compensation Loop frequency compensation is performed on the output of the error amplifier (VC pin) with a series RC network. The main pole is formed by the series capacitor and the output impedance ( 1M) of the error amplifier. The pole falls in the range of 5Hz to 30Hz. The series resistor creates a "zero" at 2kHz to 10kHz, which improves loop stability and transient response. A second capacitor, typically one tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the VC pin. VC pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. Without the second capacitor, VC pin ripple is: VC Pin Ripple = 1.245(VRIPPLE)(gm)(RC) VOUT
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LT1373
APPLICATIO S I FOR ATIO
VRIPPLE = output ripple (VP-P) gm = error amplifier transconductance ( 375mho) RC = series resistor on VC pin VOUT = DC output voltage To prevent irregular switching, VC pin ripple should be kept below 50mVP-P. Worst-case VC pin ripple occurs at maximum output load current and will also be increased if poor quality (high ESR) output capacitors are used. The addition of a 0.001F capacitor on the VC pin reduces switching frequency ripple to only a few millivolts. A low value for RC will also reduce VC pin ripple, but loop phase margin may be inadequate. Switch Node Considerations For maximum efficiency, switch rise and fall time are made as short as possible. To prevent radiation and high frequency resonance problems, proper layout of the components connected to the switch node is essential. B field (magnetic) radiation is minimized by keeping output diode, switch pin and output bypass capacitor leads as short as possible. E field radiation is kept low by minimizing the length and area of all traces connected to the switch pin. A ground plane should always be used under the switcher circuitry to prevent interplane coupling.
TYPICAL APPLICATIONS N
Positive-to-Negative Converter with Direct Feedback
VIN 2.7V TO 16V
+
T1* 2 D2 P6KE-15A D3 1N4148 1 * 4 *
C1 22F
+
5 VIN VSW LT1373 NFB 8
OFF
ON 4 S/S
3 R2 2.55k 1% R3 2.49k 1%
3
D1 MBRS130LT3
VC 1 C2 0.01F R1 5k
GND 6, 7
MAX IOUT VIN 3V 5V 9V
IOUT 0.3A 0.5A 0.75A
*COILTRONICS CTX20-2 (407) 241-7876
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The high speed switching current path is shown schematically in Figure 3. Minimum lead length in this path is essential to ensure clean switching and low EMI. The path including the switch, output diode and output capacitor is the only one containing nanosecond rise and fall times. Keep this path as short as possible.
L1 SWITCH NODE VOUT VIN HIGH FREQUENCY CIRCULATING PATH LOAD
LT1373 * F03
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Figure 3
More Help For more detailed information on switching regulator circuits, please see AN19. Linear Technology also offers a computer software program, SwitcherCADTM, to assist in designing switching converters. In addition, our applications department is always ready to lend a helping hand.
SwitcherCAD is a trademark of Linear Technology Corporation.
Dual Output Flyback Converter with Overvoltage Protection
R2 275k 1% R1 302.6k 1% VIN 4.75V TO 13V
-VOUT -5V
C3 47F
+
C1 100F
MBRS140T3 T1* 2, 3 5 + P6KE-20A * 1N4148 6, 7 *4 8 *
VOUT 15V C3 47F
2
5 VIN 8 VSW LT1373 NFB 3
OFF
FB ON 4 S/S
+
C4 47F -VOUT -15V R4 12.4k 1% R5 2.49k 1%
1
VC 1
LT1373 * TA03
GND 6, 7 C2 0.01F R3 5k
MBRS140T3
*DALE LPE-4841-100MB (605) 665-9301
LT1373 * TA04
LT1373
TYPICAL APPLICATIO S
Low Ripple 5V to - 3V "Cuk" Converter
VIN 5V L1* 2 1* 3 *4 R1 1k 1% VOUT -3V 250mA
C1 22F 10V
PACKAGE DESCRIPTION
0.300 - 0.325 (7.620 - 8.255)
0.009 - 0.015 (0.229 - 0.381)
0.065 (1.651) TYP 0.125 (3.175) 0.020 MIN (0.508) MIN 0.018 0.003 (0.457 0.076)
(
+0.035 0.325 -0.015 +0.889 8.255 -0.381
)
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
0.010 - 0.020 x 45 (0.254 - 0.508) 0.008 - 0.010 (0.203 - 0.254) 0- 8 TYP
0.014 - 0.019 (0.355 - 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
0.016 - 0.050 (0.406 - 1.270)
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of circuits as described herein will not infringe on existing patent rights.
U
U
C2 47F 16V 5
+
VIN S/S LT1373 GND GND S
VSW
8
4 7 6
+
3 1 D1** C3 47F 16V C6 0.1F
NFB VC
+
R4 5k C4 0.01F *SUMIDA CLS62-100L **MOTOROLA MBR0520LT3 PATENTS MAY APPLY R2 5.49k 1%
LT1373 * TA05
Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.045 - 0.065 (1.143 - 1.651) 0.130 0.005 (3.302 0.127) 0.400* (10.160) MAX 8 7 6 5
0.255 0.015* (6.477 0.381)
1
2
3
4
N8 1098
0.100 (2.54) BSC
S8 Package 8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 - 0.197* (4.801 - 5.004) 0.053 - 0.069 (1.346 - 1.752) 8 0.004 - 0.010 (0.101 - 0.254) 0.228 - 0.244 (5.791 - 6.197) 0.150 - 0.157** (3.810 - 3.988) 7 6 5
0.050 (1.270) BSC
SO8 1298
1
2
3
4
11
LT1373
TYPICAL APPLICATIO S
90% Efficient CCFL Supply
5mA MAX LAMP 10 T1 VIN 4.5V TO 30V 5 4 3 2 C1 0.1F 330 Q1 1N5818 2.7V TO 5.5V L1 100H 5 VIN S/S LT1373 VFB GND 6, 7 VC 2 22k 2F 1N4148 OPTIONAL REMOTE DIMMING C1 = WIMA MKP-20 L1 = COILCRAFT D03316-104 Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001 T1 = COILTRONICS CTX 110609 * = 1% FILM RESISTOR DO NOT SUBSTITUTE COMPONENTS COILTRONICS (407) 241-7876 COILCRAFT (708) 639-6400 0.1F VSW 8 562* 20k DIMMING 10k
C1 = AVX TPSD 336M020R0200 C2 = TOKIN 1E225ZY5U-C203-F C3 = AVX TPSD 107M010R0100 L1 = COILTRONICS CTX33-2, SINGLE INDUCTOR WITH TWO WINDINGS
C2 27pF
+
10F
OFF
+
2.2F 4
OFF
ON
+
1
CCFL BACKLIGHT APPLICATION CIRCUITS CONTAINED IN THIS DATA SHEET ARE COVERED BY U.S. PATENT NUMBER 5408162 AND OTHER PATENTS PENDING
RELATED PARTS
PART NUMBER LT1172 LTC 1265 LT1302 LT1308A/LT1308B LT1370 LT1372 LT1374 LT1376 LT1377 LT1613 LT1615 LT1949
(R)
DESCRIPTION 100kHz 1.25A Boost Switching Regulator 13V 1.2A Monolithic Buck Converter Micropower 2A Boost Converter 600kHz 2A Switch DC/DC Converter 500kHz High Efficiency 6A Boost Converter 500kHz 1.5A Boost Switching Regulator 4.5A, 500kHz Step-Down Converter 500kHz 1.5A Buck Switching Regulator 1MHz 1.5A Boost Switching Regulator 1.4MHz Switching Regulator in 5-Lead SOT-23 Micropower Step-Up DC/DC in 5-Lead SOT-23 600kHz, 1A Switch PWM DC/DC Converter
12
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 q FAX: (408) 434-0507 q www.linear-tech.com
U
Two Li-Ion Cells to 5V SEPIC Conveter
VIN 4V TO 9V
D1 1N4148 1
ON 4 S/S 5 VIN VSW LT1373 FB GND 6, 7 VC 1 R1 5k C4 0.01F
8
*
L1A C2 D1 33H 2.2F MBRS130LT3 * L1B 33H VOUT 5V R2 75k 1%
+
Q2
C1 33F 20V
2
+
C3 100F 10V
D2 1N4148
R3 24.9k 1%
MAX IOUT VIN 4V 5V 7V 9V
IOUT 0.45A 0.55A 0.65A 0.72A
LT1373 * TA07
LT1372 * TA06
COMMENTS Also for Flyback, Buck and Inverting Configurations Includes PMOS Switch On-Chip Converts 2V to 5V at 600mA 5V at 1A from a Single Li-Ion Cell 6A, 0.065 Internal Switch Also Regulates Negative Flyback Outputs 4.5A, 0.07 Internal Switch Handles Up to 25V Inputs Only 1MHz Integrated Switching Regulator Available 5V at 200mA from 4.4V Input 20A IQ, 36V, 350mA Switch 1.1A, 0.5, 30V Internal Switch, VIN as Low as 1.5V
1373fb LT/TP 0200 2K REV B * PRINTED IN THE USA
(c) LINEAR TECHNOLOGY CORPORATION 1995

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